System and method for a MEMS sensor

ABSTRACT

A measurement method includes generating, by a sensor, a response signal in response to an excitation signal. The method also includes generating a sampling clock signal in accordance with a pseudo-random jitter, and sampling the response signal in accordance with the sampling clock signal to determine a plurality of digital samples. The method also includes combining the plurality of digital samples to form a measurement sample.

This application claims the benefit of U.S. Provisional Application No.62/150,027, filed on Apr. 20, 2015, which application is herebyincorporated herein by reference in its entirety.

CROSS-REFERENCE TO RELATED APPLICATIONS

This patent application further relates to the following co-pending andcommonly assigned U.S. patent applications that also claim the benefitof U.S. Provisional Application No. 62/150,027, filed on Apr. 20, 2015:Ser. No. 15/074,510, filed on Mar. 18, 2016, and entitled “System andMethod for a Capacitive Sensor”, Ser. No. 15/074,649, filed on Mar. 18,2016, and entitled “System and Method for a MEMS Sensor”, and Ser. No.15/078,733, filed on Mar. 23, 2016, and entitled “System and Method fora MEMS Sensor”, which applications are hereby incorporated herein byreference in their entirety.

TECHNICAL FIELD

The present invention relates generally to a system and method formeasurement, and, in particular embodiments, to a system and method formeasurement using a sensor with pseudo-random jitter.

BACKGROUND

Microelectromechanical Systems (MEMS), which in general includeminiaturizations of various electrical and mechanical components, areproduced by a variety of materials and manufacturing methods, and areuseful in a wide variety of applications. These applications includeautomotive electronics, medical equipment, and smart portableelectronics such as cell phones, Personal Digital Assistants (PDAs),hard disk drives, computer peripherals, and wireless devices. In theseapplications, MEMS may be used as sensors, actuators, accelerometers,switches, micro-mirrors and many other devices. MEMS are also desiredfor use in environmental pressure measurement systems to measure eitherabsolute or differential environmental pressures.

When designing a system that uses a MEMS device as a sensor, variousattributes that may be taken into account include, for example,resolution and temperature sensitivity. Any ringing noise and energylosses caused by mechanical resonances of the MEMS device may also beconsidered. In some systems, such mechanical resonances may generateoscillations in response to an excitation signal, and these oscillationsmay have energy losses characterized by a Quality factor (Q). A higher Qindicates a lower rate of energy loss relative to the stored energy ofthe resonator, and thus mechanical oscillations die out more slowly. Alower Q indicates a higher rate of energy loss relative to the storedenergy of the resonator, and therefore mechanical oscillations die outmore quickly.

SUMMARY OF THE INVENTION

In accordance with a first example embodiment of the present invention,a measurement method is provided. The method includes generating, by asensor, a response signal in response to an excitation signal. Themethod also includes generating a sampling clock signal in accordancewith a pseudo-random jitter, and sampling the response signal inaccordance with the sampling clock signal to determine a plurality ofdigital samples. The method also includes combining the plurality ofdigital samples to form a measurement sample.

In accordance with a second example embodiment of the present invention,a measurement circuit is provided. The measurement circuit includes asensor. The measurement circuit is configured to generate a responsesignal in response to an excitation signal. The measurement circuit isalso configured to generate a sampling clock signal in accordance with apseudo-random jitter and sample the response signal in accordance withthe sampling clock signal to determine a plurality of digital samples.The measurement circuit is also configured to combine the plurality ofdigital samples to form a measurement sample.

In accordance with a third example embodiment of the present invention,a measurement device is provided. The measurement device includes asensor and an Analog-to-Digital Converter (ADC) coupled to an output ofthe sensor. The ADC includes a pseudo-random sequence generator and afirst oscillator. The first oscillator includes an input coupled to anoutput of the pseudo-random sequence generator. The measurement devicealso includes a filter having an input coupled to an output of the ADC.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present invention, and theadvantages thereof, reference is now made to the following descriptionstaken in conjunction with the accompanying drawings, in which:

FIG. 1 is a block diagram illustrating a pressure measurement devicethat includes a MEMS-based sensor in accordance with embodiments of thepresent invention;

FIGS. 2a thru 2 c illustrate a MEMS capacitor array layout in bridgeconfiguration and cross section of an MEMS capacitor;

FIG. 3 is a graph illustrating waveforms for a differential responsesignal having ringing noise and for a square-wave excitation signal inaccordance with embodiments of the present invention;

FIG. 4 illustrates a current embodiment input and output waveforms for aMEMS capacitor sensors arranged in a bridge configuration;

FIG. 5 illustrates an embodiment slope control circuit;

FIG. 6 illustrates an embodiment Digital Pressure Measurement systemutilizing a slope control circuit;

FIG. 7 illustrates a flow chart of an embodiment method;

FIG. 8 is a block diagram illustrating an ADC that samples adifferential response signal using pseudo-random sampling clock jitterin accordance with embodiments of the present invention;

FIG. 9 is a graph illustrating the relative ringing error at a range ofvalues for the resonant frequency of the MEMS-based sensor in accordancewith embodiments of the present invention;

FIG. 10 is a block diagram illustrating a circuit that generates avariable clock signal having a pseudo-random jitter in accordance withembodiments of the present invention;

FIG. 11 is a flow diagram illustrating a measurement method inaccordance with embodiments of the present invention;

FIGS. 12a and 12b illustrate a MEMS capacitor array schematic and layoutshowing the different dimensions and the location;

FIG. 13 is a graph illustrating output waveforms for square waveexcitation signal driving a MEMS capacitor sensors array having samesizes and resonant frequencies;

FIG. 14 is a graph illustrating output waveforms for square waveexcitation signal driving the MEMS capacitor sensors array havingdifferent sizes and resonant frequencies;

FIG. 15 is a graph illustrating frequency spectrum of the output waveform from the MEMS capacitor sensors array having different sizes andresonant frequencies;

FIG. 16; is a graph illustrating a ringing amplitude for output waveforms from the MEMS capacitor sensors array having same dimensions andresonant frequencies and MEMS capacitor sensors array having differentdimensions and resonant frequencies;

FIG. 17 illustrates a system block diagram of an embodiment MEMSpressure sensor system;

FIG. 18 illustrates a schematic block diagram of a further embodimentMEMS pressure sensor system;

FIGS. 19a and 19b illustrate waveform diagrams of example noise signalsgenerated in a sigma-delta analog to digital converter (ADC);

FIG. 20 illustrates a waveform diagram of noise signals generated in asigma-delta analog to digital converter (ADC) without a dithered clockand with a dithered clock;

FIGS. 21a and 21b illustrate schematic block diagrams of embodimentsigma-delta analog to digital converters (ADCs); and

FIG. 22 illustrates a block diagram of an embodiment method of operationfor a sensor.

Corresponding numerals and symbols in different figures generally referto corresponding parts unless otherwise indicated. The figures are drawnto clearly illustrate the relevant aspects of the preferred embodimentsand are not necessarily drawn to scale. To more clearly illustratecertain embodiments, a letter indicating variations of the samestructure, material, or process step may follow a figure number.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

The making and using of the presently preferred embodiments arediscussed in detail below. It should be appreciated, however, that thepresent invention provides many applicable inventive concepts that canbe embodied in a wide variety of specific contexts. The specificembodiments discussed are merely illustrative of specific ways to makeand use the invention, and do not limit the scope of the invention.

The present invention will be described with respect to preferredembodiments in a specific context, a system and method for performing ameasurement a capacitive pres sure-measurement system that uses aMEMS-based sensor. Further embodiments may be applied to other sensorsystems such as, for example, piezo-resistive sensor systems. Some ofthe various embodiments described herein include capacitive MEMSpressure sensors, interface circuits, sigma-delta analog to digitalconverters (ADCs) for MEMS pressure sensor interface circuits, noise ininterface circuits, and dithered clocks for sigma-delta ADCs andinterface circuits. In other embodiments, aspects may also be applied toother applications involving any type of transducer system according toany fashion as known in the art.

A capacitive MEMS pressure transducer uses a pressure difference betweentwo regions to adjust a variable capacitance structure and generate anoutput signal proportional to the pressure difference. In one specificapplication, a differential output capacitive MEMS pressure transduceruses two variable capacitance structures to generate a differentialoutput that varies according to the measured pressures. In variousembodiments, the signal output from the pressure transducer is an analogsignal. The analog signal may be amplified and converted to a digitalsignal.

In embodiments of the present invention, a capacitive MEMS pressuresensor is operated by introducing a periodic excitation signal at afirst port of the MEMS pressure sensor, and monitoring an output of theMEMS pressure sensor at a second port of the MEMS pressure sensor. Apressure measurement is then made by determining amplitude of the signalat the second port of the MEMS sensor. One issue faced in such systemsis an underdamped response of the MEMS-based sensor due to mechanicalresonances within the MEMS-based sensor, which can lead to measurementerror in some circumstances due to the ringing nature of the outputsignal. In various embodiments, disclosed herein, systems and methods ofmeasuring such an underdamped system are disclosed.

In a first embodiment, in order to reduce error caused by theunderdamped response of the MEMS pressure sensor to the excitationsignal, the slope of the excitation signal is reduced in order toattenuate harmonics that may stimulate the underdamped response of theMEMS pressure sensor. In some embodiments, the excitation signal isconditioned or generated such that instances of sharp edges are reducedor eliminated. In a specific embodiment, slope reduction is achieved bygenerating a dual slope integrated triangular waveform from a squarewave input signal and integrating this signal again to generate aperiodic waveform having a smooth transition between a first signallevel and a second signal level. In some embodiments, the slope of thefirst square wave signal is controlled by delay-locked loop in order tosynchronize an edge of the slope reduced excitation pulse with anincoming clock signal.

In a second embodiment, the output of the MEMS-based sensor is measuredusing a dithered sampling clock. By dithering the sampling time withrespect to the underdamped response of the MEMS, a series ofmeasurements may be taken in which the underdamped component of the MEMSresponse averages out. More specifically, a period of a variable clocksignal is changed at regular intervals that correspond to a switchingfrequency. At a switching time after each such regular interval, theperiod of the variable clock signal is increased relative to a minimumperiod by a period adjustment amount that is pseudo-randomly determined.Equivalently, the frequency of the variable clock signal is decreased bya frequency shift that corresponds to the period adjustment amount. Theswitching frequency of the variable clock signal is designed to be closeto the mechanical resonant frequency of a MEMS-based sensor. A samplingclock signal is derived by dividing the frequency of the variable clocksignal. To spread the resonant ringing noise of the MEMS-based sensoroutput, this output is digitally sampled at pseudo-randomly varyingintervals in accordance with the sampling clock signal. Multiple digitalsamples are then filtered and combined to suppress the wideband ringingnoise.

In a third embodiment, the MEMS pressure sensor is implemented using anarray or MEMS pressure sensors that have varying dimensions, such thateach of the MEMS pressure sensors resonate at different frequencies.Hence, when the MEMS pressure sensors are stimulated by the excitationsignal, the amplitude of the ringing may be reduced at various times dueto the various resonant responses being out of phase with each other. Bysampling the output of the MEMS sensor when the destructive interferenceof the various MEMS pressures reduces the amplitude of ringing response,a more accurate measurement may be made.

In a fourth embodiment, an oversampled analog to digital converter (ADC)is used to monitor the output of the MEMS sensor. In order to alleviateidle tones in the oversampled ADC, a dithered clock is used to operatethe oversampled ADC. In some embodiments, the dither clock signal may begenerated according to the second embodiment disclosed herein.

FIG. 1 shows an embodiment pressure measurement device 100 that includesa sensor 103. The sensor 103 is coupled to outputs of an excitationsignal generator 102. The excitation signal generator 102 generates analternating excitation signal that it provides to the sensor 103, whichgenerates an analog measurement signal made up of two excitationresponse signals. Each of these excitation response signals is generatedby oscillations of a resonance of sensor 103 that may be, e.g., amechanical resonance. In an embodiment, the sensor 103 has anunderdamped response.

Referring again to FIG. 1, the sensor 103 includes a capacitance bridgehaving two bridge sections 105. Each bridge section 105 respectivelyincludes a pressure-sensitive capacitor 111 in series with a referencecapacitor 109, and one of the response signals of the sensor 103 isoutput at a center tap between the pressure-sensitive capacitor 111 andthe reference capacitor 109. The reference capacitors 109 have acapacitance of C_(r) that is relatively stable over pressure as comparedto the capacitance C_(s) of the pressure-sensitive capacitors 111. In anembodiment, the pressure-sensitive capacitors 111 are respectivelyimplemented using one or more high-Q MEMS elements included in thesensor 103, and these MEMS elements have a mechanical resonance with aresonant frequency f_(r). In some embodiments, the reference capacitors109 are implemented using capacitors selected such that C_(r) changeswith temperature in a known relationship relative to thetemperature-induced change of C_(s).

A readout amplifier 104 coupled to outputs of the sensor 103 amplifiesthese sensor response signals. An analog-to-digital converter (ADC) 106coupled to outputs of the readout amplifier 104 then samples thedifference between the amplified sensor response signals to providedigital samples. A filter 108 coupled to the ADC 106 combines several ofthese digital samples over a time interval to generate a single pressuremeasurement sample. In some embodiments, the filter 108 is a low-passfilter that averages the digital samples. In other embodiments, thefilter 108 combines the digital samples using a more complex algorithm,which may include, for example, selecting the sample having the medianvalue, discarding outlier samples before averaging, etc.

In some embodiments, each of the excitation signal generator 102, thesensor 103, the readout amplifier 104, the ADC 106, and the filter 108of the pressure measurement device 100 are included in a singleIntegrated Circuit (IC), and this IC has a volume that is less than 10cubic millimeters. In other embodiments, multiple ICs may be included inthe pressure measurement device 100.

FIGS. 2a-c illustrate an example implementation of a MEMS sensor thatmay be used to implement sensor 103 shown in FIG. 1. As shown, in FIG.2a , the MEMS sensor is arranged in a bridge configuration that includesfixed capacitors C_(r) and variable capacitors C_(s). In an embodimentvariable capacitors C_(s) are each implemented using an array of MEMSsensors, while the fixed capacitors are implemented using an array offixed capacitors that are designed to track a nominal capacitance of thearray of MEMS sensors.

FIG. 2b shows an example embodiment layout 210 of the MEMS capacitorsensor array 200. As shown, the layout includes fixed capacitance cells212 and 218 that are used to implement capacitances C_(r), and MEMSsensor cells 214 and 216 that are sensitive to pressure. As shown,layout 210 configuration may be arranged to avoid gradient mismatchbetween all four capacitors in the bridge. In an embodiment, MEMS sensorcells 214 may be implemented using MEMS sensor structures known in theart, while fixed capacitance cells may be implemented using MEMS sensorcells whose motion is disabled. By using a similar physical structurefor both MEMS sensor cells 214 and 216 and fixed capacitance cells 212and 218, good matching between MEMS sensor cells 214 and 216 and fixedcapacitance cells 212 and 218 may be achieved over process andtemperature variations. In some embodiments, the motion of fixedcapacitance cells 212 and 218 may be prevented, for example, by notopening a pressure port during processing of the semiconductorcomponent, or by adding mechanical motion barriers within the MEMSstructure.

FIG. 2c illustrates a cross-section of a MEMS sensor cell 220 alongsidean example NMOS and PMOS transistor. As shown, the MEMS sensor cell isformed with a poly-silicon membrane acting as a top electrode 222 and afixed counter electrode 224 to form the sensor cell. There is a vacuumcavity 226 between the top electrode and the fixed counter electrode.The top electrode 222 is not covered to allow any pressure applicationand causing a change in capacitor value. It should be appreciated thatcross-section 220 is just one of many examples of suitable MEMS cellsthat may be used in embodiments of the present invention.

FIG. 3 shows waveforms for an embodiment square-wave excitation signal302 and a differential response signal 304. The square-wave excitationsignal is one embodiment of an alternating excitation signal that may beproduced by the excitation signal generator 102 of FIG. 1. In otherembodiments, any alternating excitation signal may be used, includingfor example, a sinusoidal signal, a triangular signal, a compositesignal, etc.

Referring again to FIG. 3, the differential response signal 304represents the difference between the response signals at the outputs ofthe sensor 103 of FIG. 1. Because the one or more MEMS elements thatmake up the pressure-sensitive capacitors 111 are high Q, the mechanicalresonance of these MEMS elements is under-damped and mechanicaloscillations die out slowly. At various sampling times t1, t2, and t3, aringing noise is therefore introduced into the differential responsesignal 304 relative to the ideal response signal 306 of a sensor formedfrom ideal capacitors.

First Embodiment

In the first embodiment, the slope of the excitation 302 is reduced inorder to avoid overly stimulating the resonant condition of MEMS sensor103. FIG. 4 illustrates a comparison between a square wave excitationsignal 402, and wave shaped excitation signal 404 that are used tostimulate MEMS sensor at input port Vex. As shown, output waveform 306represents high amplitude ringing due to a square wave input 402 thatcontains steep slope at rising and falling edges because of theresonances stimulated by square wave excitation signal 402 having steeprising and falling edges causes resonance. The output waveform 408, onthe other hand, represents the response from the wave shaped excitationsignal 404 and exhibits very little ringing due to smooth edges at thefalling and rising portion of the input as well as the transition regionto the flat region. The smooth edges at the rising and falling portionof the excitation signal 404 reduces the effect of resonances due tohigh Q Factor of the MEMS sensors and capacitors and provide a moresmoother output waveform 408.

In an embodiment, the slope of the excitation signal is controlled inthe time domain by using cascaded integrators to control the rising andfalling behavior of the excitation signal, while maintaining a flatoutput voltage between edge transitions. Accordingly, the firstintegration produces a triangular edge, while the second integrationproduces an edge having a second order or parabolic shape. The output ofthe second integration is used to drive a MEMS sensor 103 arranged in abridge configuration. Such an embodiment shaping reduces the amplitudeof the generated harmonics and reduces the ringing seen at the output ofMEMS sensor 103. It should be understood that the cascaded integratorapproach is just one example of many possible embodiment systems andmethods that may be used to control the slope of the excitation signal.In some embodiments, the excitation signal that has a time period inwhich the signal value is stable having, for example, a fixed referencevoltage. This time period in which the time period is stable may also bereferred to as a “flat region.” Within this time period, the sensoroutput signal and readout amplifier output signal will also be stableand can be sampled, for example, by ADC 106 shown in FIG. 1.

FIG. 5 illustrates an embodiment excitation pulse generation system 500that includes a timing control circuit 502, a charge pump 504, a firstintegrating capacitor C1, a second integrator and wave shaper circuit508, a switch 510 for controlling a second integrator output to becoupled to C_(Load), a comparator 512, a phase detector 514, and secondcharge pump 516 driving a loop filter capacitor C2. The timing controlcircuit 502 accepts the input square wave signal or clock from pulsegenerator 501 and generates two switch control signals for driving thecharge pump 504. These two switch control signals selectively activateswitches that connect a charging or discharging current source to theintegrating capacitor C1. The square wave clock signal is integratedacross the integrating capacitor C1 and an integrated triangularwaveform is generated. This integrated triangular waveform is bufferedby the buffer amplifier 506. The buffered triangular waveform is furtherintegrated to generate a wave shape that contains very low energycontent at the resonant frequency of the MEMS capacitor. The secondintegration smoothens out the sharp edges present in a square waveformand the edge of the triangular wave form. It is understood that a squarewave having sharp edges contain a lot of high frequency components thatcan stimulate a resonant response within MEMS sensor 103. The endpointsof the integrated wave shape is controlled to ensure that edges of theslope controlled output signal is synchronized with the input clocksignal. A switch 510 is configured to couple the load capacitance CLoadto either the output of integrator and wave shaper circuit 508 or supplyvoltage VDD of this circuit block (which typically is a temperaturestable and low noise reference voltage). In some embodiments, switch 510is used to couple the load capacitance C_(Load) to supply voltage VDD incase the output of second integrator and wave shaper 508 does not meetthe supply voltage.

As shown in FIG. 5, the sloped controlled excitation signal runs thoughcomparator 512 to form a signal that may be used to adjust the phasedifference between input clock and the excitation signal. The phase ofthe signal at the output of comparator 512 is compared with the phase ofthe input clock signal via phase detector 514, which generates twocontrol signals to activate two switches in a charge pump circuit 516.The switches connect a charging and a discharging current source to aloop filter capacitor C2. The voltage at the loop filter capacitor C2 isan indication of the phase difference between the excitation signal andthe input clock signal. The loop filter capacitor C2 transforms aninstantaneous phase difference to an analog voltage. This voltage isused to control the amplitude of the charging and discharging currentwhile generating an integrated triangular waveform. In an embodiment,the excitation pulse generation system 500 can be implemented in asingle integrated circuit (IC).

FIG. 6 illustrates an embodiment Digital Pressure Measurement System 600that includes embodiment excitation signal generator 602, a capacitivepressure sensor 604 (including the readout amplifier 104), a temperaturesensor 606, a mux 608, an analog to digital converter (ADC) 610, adigital signal processing 612, a digital core 614, a digital interface616, a voltage regulators 618, a memory interface 620, a unit storingcalibration coefficients 622 and FIFO (First In First Out) 624. Theexcitation signal generator 602 provides the slope controlled excitationsignal to the capacitive pressure sensor 604 according to embodimentsdescribed above. The mux 608 selects measurement from temperature sensor606 or capacitor sensor 604 and sends to an ADC circuit 610 for digitalconversion of the measurements. The ADC output is then passed through adigital signal processing unit 612 for further filtering andmathematical computation. The digital core 614 and the digital interface616 are part of the internal processor that converts the temperature andpressure measurements into 24 bits of digital word. The calibrationco-efficient 622 stores calibrated values for each individual pressuresensors to be used for measurement correction. The FIFO 624 storesmultiple temperature and pressure measurements during low power mode.The memory interface 620 provides these values to digital core 614. Theembodiment also includes an internal voltage regulator for supplyingpower to internal circuits.

In an embodiment, Digital Pressure Measurement System 600 may beimplemented using a single integrated circuit and/or a combination ofintegrated circuits and/or discreet components. It should be appreciatedthat system 600 is just one of many example systems in which anembodiment excitation signal generator may be implemented.

FIG. 7 illustrates a flowchart of an embodiment method 700 ofcontrolling slope of a MEMS capacitor. In step 702, a first integrationis performed on a first input signal. In an embodiment the first inputsignal is a square wave signal. Next in step 704, a second integrationis performed on the first integration output signal. In an embodiment,the first integration output is a triangular waveform and the secondintegration is performed for a wave shaping. When the output signal hasreached the reference voltage (Vdd) or a certain amount time has passed,the output is kept constant (eventually the output is switched to Vdd)until a falling edge procedure is triggered in some embodiments. Next instep 706, the output of the second integration is used to drive a MEMScapacitor bridge, where bridge is having two sections and each sectionis comprised of one pressure sensitive capacitor and one referencecapacitor. Next in step 708, the phase of the input signal and output ofthe second integration are synchronized. In an embodiment, a phasedetector is used to synchronize the phases. Next in step 710, a readoutamplifier connected to the common point of the pressure sensitivecapacitor and the reference capacitor is used to measure instantaneouscapacitor changes. Finally in step 712, an A/D conversion is performedon the output of the readout amplifier to calculate the pressure.

In accordance with various embodiments, circuits or systems may beconfigured to perform particular operations or actions by virtue ofhaving hardware, software, firmware, or a combination of them installedon the system that in operation causes or cause the system to performthe actions. One general aspect includes a method of performing ameasurement with a capacitive sensor, the method including: generating aperiodic excitation signal, the periodic excitation signal including aseries of pulses; smoothing edge transitions of the series of pulses toform a shaped periodic excitation signal; providing the shaped periodicexcitation signal to a first port of the capacitive sensor; andmeasuring a signal provided by a second port of the capacitive sensor.Other embodiments of this aspect include corresponding circuits andsystems configured to perform the various actions of the methods.

Implementations may include one or more of the following features. Themethod further including determining an output measurement based on themeasured signal. The method where the determined output measurementincludes a pressure measurement. The method where smoothing edgetransitions includes generating a first sloped signal based on thegenerated periodic excitation signal to form a sloped excitation signal.The method where smoothing edge transitions further includes integratingthe sloped signal to form the shaped periodic excitation signal. Themethod further including adjusting a slope of the first sloped signalbased on a timing difference between the shaped periodic excitation andthe periodic excitation signal. The method where generating the firstsloped signal includes charging a capacitor with a first current sourceand discharging the capacitor with a second current source. The methodfurther including adjusting a slope of the first sloped signal based ona timing difference between the shaped periodic excitation and theperiodic excitation signal, where adjusting the slope includes adjustinga current of the first current source and the second current source. Themethod further including determining the timing difference between theshaped periodic excitation signal and the periodic excitation signal.The method where determining the timing difference includes using aphase detector. The method where the capacitive sensor includes a MEMSsensor. The method where the MEMS sensor includes a sensor bridge havinga first branch having a first MEMS pressure sensor and a firstcapacitor, and a second branch having a second MEMS pressure sensor anda second capacitor. Implementations of the described techniques mayinclude hardware, a method or process, or computer software on acomputer-accessible medium.

A further general aspect includes a system including: an excitationgenerator configured to be coupled to a first port of a capacitivesensor, the excitation generator including a pulse generator, and apulse smoothing circuit coupled to an output of the pulse generator,where an output of the pulse smoothing circuit is configured to becoupled to the first port of the capacitive sensor. Other embodiments ofthis aspect include corresponding circuits and systems configured toperform the various actions of the methods.

Implementations may include one or more of the following features. Thesystem further including a readout circuit configured to be coupled to asecond port of the capacitive sensor. The system where the readoutcircuit includes an A/D converter configured to be coupled to the secondport of the capacitive sensor. The system where the readout circuit isconfigured to determine a response of the capacitive sensor based on asignal emanating from the second port of the capacitive sensor. Thesystem further including the capacitive sensor. The system where thecapacitive sensor includes a MEMS sensor. The system where the MEMSsensor includes a sensor bridge having a first branch having a firstMEMS pressure sensor and a first capacitor, and a second branch having asecond MEMS pressure sensor and a second capacitor. The system where thepulse smoothing circuit includes a ramp generator having an inputcoupled to the output of the pulse generator. The system where the rampgenerator includes a first current source and a second current sourcecoupled to a first capacitor. The system where the pulse smoothingcircuit further includes an integrator coupled to an output of the rampgenerator, where an output of the integrator is coupled to the output ofthe pulse smoothing circuit. The system further including a phasedetector having a first input coupled to the output of the pulsegenerator and a second input coupled to the output of the integrator,where an output of the phase detector is configured to control a slopeof a signal at the output of the ramp generator. The system furtherincluding a charge pump coupled to an output of the phase detector, anda second capacitor coupled to an output of the charge pump. The systemwhere the slope of the signal at the output of the ramp generator isbased on a voltage across the second capacitor. The system where theexcitation generator is disposed on an integrated circuit. The systemwhere the capacitive sensor is further disposed on the integratedcircuit. The system where the pulse smoothing circuit includes: a firstintegrator coupled to the output of the pulse generator; and a secondintegrator coupled to an output of the first integrator, where an outputof the second integrator is coupled to an output of the pulse smoothingcircuit. The system further including a phase detector having a firstinput coupled to the output of the pulse generator and a second inputcoupled to the output of the integrator, where an output of the phasedetector is configured to control a slope of the first integrator. Thesystem where the first integrator includes a plurality of currentsources coupled to an integrating capacitor, and controlling the slopeof the first integrator includes adjusting a current of the plurality ofcurrent sources based on the output of the phase detector.Implementations of the described techniques may include hardware, amethod or process, or computer software on a computer-accessible medium.

An advantage of some implementations of the first embodiment includesthe ability to reduce the effect of ringing when capacitive MEMS isenabled via an excitation signal. The amount of ringing is a function ofthe excitation signal wave shape and MEMS capacitor resonant frequency.

Second Embodiment

In the second embodiment, ADC 106 varies its internal sampling clocksignal by a pseudo-random jitter to mitigate the effect of ringingnoise. This pseudo-random jitter is provided by varying the timing ofthe rising and/or falling edge of the sampling clock on which ADC 106derives its timing reference. Accordingly, the systematic ringing errorof the outputs of the sensor 103 is thus converted to a wide-bandsignal, which the filter 108 may suppress by, for example, averagingmany digital samples to form each combined measurement sample.

FIG. 8 illustrates an example ADC 800 that may be used as the ADC 106 ofFIG. 1 to generate a clock signal having pseudo-random jitter. ADC 800includes a variable clock generator 804, a frequency divider 806, and asampling unit 808.

The variable clock generator 804 generates a variable clock signal thathas a pseudo-random jitter. The variable clock generator 804 includesthis pseudo-random jitter in the variable clock signal by switching thelength of the period T_(per) of the variable clock signal. T_(per) isequal to a minimum period T_(per) _(_) _(min) increased by a periodadjustment ΔT_(per) that is switched every clock cycle of T_(per). Thesampling unit 808 receives a lower clock frequency through the clockfrequency divider 806. For example, in the embodiment of FIG. 8, theclock frequency is an integer N times lower such that the effectivesampling clock has a clock jitter of duration T_(sw) that is equal to Ntimes ΔT_(per). In an embodiment, T_(sw) is chosen such that it randomlysamples the MEMS signal having the resonance behavior at time pointsthat are spread over at least one period 1/f_(res), where f_(res) is theresonant frequency of the MEMS sensor and the resonance period T_(res)is the reciprocal of f_(res). As an example, for a situation withf_(res)=5 MHz, T_(res)=200 ns, and N=8, ΔT_(per) is made to be less than200 ns (preferably less than 200 ns/4) yet still high enough such thatT_(sw) is greater than 200 ns. In alternative embodiments, the clockfrequency divider 806 may have another division ratio and/or the MEMSsensor may have a different resonant frequency.

The variable clock generator 804 is controlled in a feedback loop by aclock controller 802 to stabilize the average period T_(per) _(_) _(avg)of the variable clock signal. The clock controller 802 is provided areference oscillator signal by a reference oscillator 803, which may bean oscillating crystal or any other form of stable electronicoscillator. In an embodiment, the clock controller 802 may include aphase-lock loop. In an embodiment, the clock controller 802 may providethe variable clock generator 804 a clock signal having a period that isdifferent from the period of T_(per) _(_) _(min), which is then scaledin frequency by the variable clock generator 804. The variable clockgenerator 804 provides a clock feedback signal to the clock controller802.

The frequency divider 806 is coupled to an output of the variable clockgenerator 804 and generates a sampling clock signal having a samplingperiod T_(sam) that is N times the period T_(per). The sampling clocksignal therefore also includes a pseudo-random jitter.

An input of the sampling unit 808 is coupled to the output of thefrequency divider 806 to receive the sampling clock signal. The samplingunit 808 also has inputs that receive the two amplified sensor responsesignals that are output from the readout amplifier 104 of FIG. 1. Thesampling unit 808 generates samples by sampling a difference betweenthese sensor response signals, and this sampling is performed everyT_(sam) seconds. These samples are quantized in one or more subsequentstages of the ADC (not shown) In an embodiment, the ADC is a sigma-deltaconverter, and the quantization stage(s) also include an additional loopfilter that filters the output of the sampling unit.

FIG. 9 shows a graph plotting the relative ringing error at an output ofthe filter 108 of FIG. 1. The maximum period shift of the sampling clockis 150 nanoseconds, 16,384 digital samples are averaged to form eachmeasurement sample, and the switching frequency f_(sw) is 5120 kHz. Theringing error is calculated relative to a resonance-free response signalof a sensor formed from ideal capacitors. This relative ringing error isminimized when the resonant frequency is equal to the switchingfrequency, and increases as the resonant frequency is varied away fromthe switching frequency in either direction.

FIG. 10 shows a block diagram of an embodiment variable clock generator804. The variable clock generator 804 includes a counter 1002, ade-multiplexer 1004, an LFSR 1006, and an oscillator 1008.

The counter 1002 has a counter reset input that receives a clock controlsignal from the clock controller 802. In the exemplary embodiment ofFIG. 10, the frequency of this clock control signal is 160 kHz. Thecounter 1002 also has a counter clock input coupled to an output of theoscillator 1008 to receive a variable clock signal. In the embodiment ofFIG. 10, the counter 1002 is a 3-bit counter that generates a countersignal that represents a count value incremented from 0 to 7synchronously with the variable clock signal received from oscillator1008, and the count value is reset to 0 by a rising edge on the counterreset input. The most significant bit of the counter signal is providedas the feedback control signal to the clock controller 802. Thisfeedback control signal is one-eighth the frequency of the variableclock signal.

The de-multiplexer 1004 also has an input coupled to the output ofcounter 1002 to receive the counter signal. The de-multiplexer 1004switches on or off a binary LFSR enable signal based on the count value.The de-multiplexer 1004 also switches on or off a first binary input toan AND gate 1010 based on the value of the counter signal.

In the example embodiment of FIG. 10, when the count value is 0 or 1,the de-multiplexer switches on the LFSR enable signal, and otherwise thede-multiplexer 1004 switches the LFSR enable signal off. When the countvalue is 3 or 4, the de-multiplexer switches on the first input to theAND gate 1010, and otherwise the de-multiplexer 1004 switches the ANDgate 1010 off. Since the count value can take any one of eight possiblevalues, the LFSR enable signal is therefore switched on during only thefirst quarter of the variable clock period of oscillator 1008, and thefirst input to the AND gate 1010 is switched on only during the nextquarter of the oscillator clock period.

The LFSR 1006 includes an enable input that receives the LFSR enablesignal from the de-multiplexer 1004. The LFSR 1006 also has a resetinput that receives the clock control signal from clock controller 802as an LFSR reset signal. The LFSR 1006 also includes a clock input thatreceives the variable clock signal that is output from the oscillator1008. Based on the LFSR enable signal and the LFSR reset signal, theLFSR 1006 generates a pseudo-random sequence that is synchronous withthe variable clock signal. In some embodiments, the LFSR 1006 is aFibonacci LFSR. In other embodiments, the LFSR 1006 is a Galois LFSR. Instill other embodiments, any pseudo-random sequence generator known inthe art, including a non-linear feedback shift register, may be used inplace of the LFSR 1006.

In the embodiment of FIG. 10, the LFSR 1006 is a 17-bit LFSR thatoutputs a LFSR state signal representing a two-bit binary state of theLFSR 1006. The LFSR 1006 provides this LFSR state signal bit-by-bit to asecond binary input of the AND gate 1010. Based on this LFSR statesignal and the signal at the first input of the AND gate 1010, the ANDgate 1010 generates a frequency select signal that is also a two-bitbinary sequence.

The clock generator 804 also includes a D flip-flop 1012 that receivesthis frequency select signal from AND gate 1010, and also receives thevariable clock signal that is output from the oscillator 1008. The Dflip-flop 1012 also has an output coupled to an input of the oscillator1008, and the D flip-flop 1012 provides the frequency select signal tothe oscillator 1008, bit-by-bit, synchronously with the variable clocksignal.

The oscillator 1008 generates the variable clock signal, which theoscillator 1008 varies based on the frequency select signal that isprovided by the D flip-flop 1012. The oscillator 1008 has an oscillatingfrequency f_(osc), the maximum of which is the reciprocal of T_(per)_(_) _(min) (shown in FIG. 8). Based on the two-bit binary value of thefrequency select signal that is received by the oscillator every twoperiods of the variable clock signal (i.e., every two oscillations), theoscillator 1008 either maintains its oscillating frequency f_(osc) atits previous value or reduces its oscillating frequency by a frequencyshift. Decreasing the frequency of the oscillator 1008 by this frequencyshift corresponds to adding a period adjustment of ΔT_(per) to T_(per)_(_) _(min) to obtain the period T_(per) of the variable clock signal.

In the embodiment of FIG. 10, the oscillator 1008 changes its frequencyshift, at each switching time, by an amount that is a reciprocal of 50nanoseconds, and the total frequency shift at any time is the reciprocalof a ΔT_(per) of either 0, 50, 100, or 150 nanoseconds. The maximumfrequency f_(osc) of the oscillator is equal to 1280 kHz, whichcorresponds to a minimum period of the variable clock signal of 781.25nanoseconds.

In an example, the frequency of the variable clock generator 804 iseight times the frequency of the sampling clock (i.e., N=8), and thus afrequency shift is determined four times during each period T_(sam) ofthe sampling clock. In this case, the oscillator 1008 may change itsfrequency shift from its previous value at each of four switching timesduring each period T_(sam) of the sampling clock signal. In thisexample, the minimum of the sampling clock period T_(sam) is eight timesthe minimum period of the variable clock signal of 781.25 nanoseconds,which is 6250 nanoseconds. This minimum duration of T_(sam) correspondsto a maximum sampling clock frequency of 160 kHz in this example. When afrequency shift corresponding to a maximum ΔT_(per) of 150 nanosecondsis applied, the oscillator has a maximum period T_(per) of 931.25nanoseconds. Since the maximum of T_(sam) is eight times this maximumT_(per), or 7450 nanoseconds, which corresponds to a maximum samplingclock frequency of 134.2 kHz in this example. The expected mean ofΔT_(per) applied over a long time interval will be 75 nanoseconds. Thisexpected mean is determined by averaging 0, 50, 100, and 150nanoseconds, which are the pseudo-randomly selected values of ΔT_(per).The expected mean of T_(per) is thus 856.25 nanoseconds, whichcorresponds in this example to a mean sampling clock period T_(sam) of6850 nanoseconds and a mean sampling frequency of 146.0 kHz.

FIG. 11 is a flow diagram illustrating an embodiment measurement method.The method begins at step 1102. At step 1104, sensor 103 generatesresponse signals in response to an excitation signal. At step 1106, acounter signal is incremented synchronously with a variable clocksignal. At step 1108, an LFSR enable signal is determined in accordancewith count values of the counter signal. At step 1110, an m-bit LFSRstate signal is determined synchronously with the variable clock signalin accordance with the LFSR enable signal and a clock control signal. Atstep 1112, a frequency select signal is determined in accordance withthe LFSR state signal and a count value of the counter signal.

At step 1114, a flow decision is made based on whether a new frequencyshift that is different from the previous frequency shift has beenselected by the frequency select signal. If a new frequency shift hasbeen selected, flow continues at step 1118, where the frequency of thevariable clock signal is switched in accordance with the selectedfrequency shift. Otherwise, flow continues at step 1116, where the lastfrequency of the variable clock signal is maintained. Flow thencontinues in either case at step 1120, where the variable clock signalis downscaled in frequency by a factor of N to obtain a sampling clocksignal. At step 1122, the response signals generated by sensor 103 aresampled in accordance with the sampling clock signal.

At step 1124, a flow decision is made based on whether enough sampleshave been collected to perform an averaging operation. This requisitenumber of samples may be based on a design setting, for example. If notenough samples have been collected, flow continues at step 1125, whereanother flow decision is made based on whether the clock control signalhas a rising edge. If a rising edge is not detected, flow continues atstep 1106.

If a rising edge is detected at step 1125, flow continues at step 1127,where the counter is reset to 0. Flow then continues at step 1108.

If enough samples for averaging have been collected at step 1124, flowcontinues at step 1126, where these samples are averaged together toobtain a combined pressure measurement sample. The method then ends atstep 1128.

In accordance with various embodiments, circuits or systems may beconfigured to perform particular operations or actions by virtue ofhaving hardware, software, firmware, or a combination of them installedon the system that in operation causes or cause the system to performthe actions. In accordance with an example embodiment of the presentinvention, a measurement method is provided. The method includesgenerating, by a sensor, a response signal in response to an excitationsignal. The method also includes generating a sampling clock signal inaccordance with a pseudo-random jitter, and sampling the response signalin accordance with the sampling clock signal to determine a plurality ofdigital samples. The method also includes combining the plurality ofdigital samples to form a measurement sample.

Also, the foregoing measurement method example embodiment may beimplemented to include one or more of the following additional features.The method may also be implemented such that generating the samplingclock signal includes generating a variable clock signal having avariable clock frequency that switches in accordance with a switchingfrequency. In this implementation, a period of the sampling clock signalis an integer multiple of a period of the variable clock signal.

The method may also be implemented such that the sensor has anunderdamped response. In this implementation, the switching frequency isnot less than 0.9 times a mechanical resonant frequency of the sensor,and the switching frequency is not greater than 1.1 times the mechanicalresonant frequency of the sensor. The method may also be implementedsuch that generating the variable clock signal includes generating, by aLinear Feedback Shift Register (LFSR), an LFSR state signal inaccordance with the variable clock signal and a reference oscillatorsignal.

The method may also be implemented such that generating the variableclock signal further includes generating a counter signal in accordancewith the variable clock signal and the reference oscillator signal, andgenerating an LFSR enable signal in accordance with the counter signal.In this implementation, the method also includes generating a frequencyselect signal in accordance with the LFSR state signal and the countersignal. Generating the LFSR state signal is further in accordance withthe LFSR enable signal, and generating the variable clock signal isfurther in accordance with the frequency select signal.

The method may also be implemented such that the excitation signalincludes a square wave, and the sensor includes aMicro-Electro-Mechanical System (MEMS) element. The MEMS element mayinclude a first pressure-sensitive capacitor.

The method may also be implemented such that the sensor further includesa capacitance bridge, and the capacitance bridge includes a pair ofbridge sections, each respectively having a pressure-sensitive capacitorand a reference capacitor coupled together at an output node. In thisimplementation, a capacitance of the pressure-sensitive capacitor varieswith pressure more than a capacitance of the reference capacitor, andthe response signal includes output signals of the pair of bridgesections. The method may also be implemented such that the capacitanceof the reference capacitor changes with temperature in a knownrelationship relative to a change with temperature of the capacitance ofthe pressure-sensitive capacitor. The method may also be implementedsuch that the combining the plurality of digital samples includes atleast one of averaging first digital samples of the plurality of digitalsamples or calculating a median value of the first digital samples.

In accordance with another example embodiment of the present invention,a measurement circuit is provided. The measurement circuit includes asensor. The measurement circuit is configured to generate a responsesignal in response to an excitation signal. The measurement circuit isalso configured to generate a sampling clock signal in accordance with apseudo-random jitter and sample the response signal in accordance withthe sampling clock signal to determine a plurality of digital samples.The measurement circuit is also configured to combine the plurality ofdigital samples to form a measurement sample.

Also, the foregoing measurement circuit example embodiment may beimplemented to include one or more of the following additional features.The measurement circuit may also be further configured to generate avariable clock signal having a variable clock frequency that switches inaccordance with a switching frequency. In this implementation, a periodof the sampling clock signal is an integer multiple of a period of thevariable clock signal. The measurement circuit may also be configuredsuch that the sensor has an underdamped response. In thisimplementation, the switching frequency is not less than 0.9 times amechanical resonant frequency of the sensor, and the switching frequencyis not greater than 1.1 times the mechanical resonant frequency of thesensor. The measurement circuit may also be implemented to furtherinclude an LFSR configured to generate an LFSR state signal inaccordance with the variable clock signal and a reference oscillatorsignal.

The measurement circuit may also be further configured to generate acounter signal in accordance with the variable clock signal and thereference oscillator signal. In this implementation, the measurementcircuit may also be configured to generate the variable clock signal inaccordance with a frequency select signal, generate an LFSR enablesignal in accordance with the counter signal, and generate the frequencyselect signal in accordance with the LFSR state signal and the countersignal. The LFSR may also be further configured to generate the LFSRstate signal in accordance with the LFSR enable signal.

The measurement circuit may also be implemented such that the excitationsignal includes a square wave, and the sensor includes a MEMS element.In this implementation, the MEMS element may include a firstpressure-sensitive capacitor.

The measurement circuit may also be implemented such that the sensorfurther includes a capacitance bridge. The capacitance bridge may alsoinclude a pair of bridge sections, each respectively including apressure-sensitive capacitor and a reference capacitor coupled togetherat an output node. In this implementation, a capacitance of thepressure-sensitive capacitor varies with pressure more than acapacitance of the reference capacitor, and the response signal includesoutput signals of the pair of bridge sections. The measurement circuitmay also be implemented such that the capacitance of the referencecapacitor changes with temperature in a known relationship relative to achange with temperature of the capacitance of the pressure-sensitivecapacitor. The measurement circuit may also be implemented such that themeasurement sample includes at least one of an average of first digitalsamples of the plurality of digital samples or a median value of thefirst digital samples.

In accordance with another example embodiment of the present invention,a measurement device is provided. The measurement device includes asensor and an Analog-to-Digital Converter (ADC) coupled to an output ofthe sensor. The ADC includes a pseudo-random sequence generator and afirst oscillator. The first oscillator includes an input coupled to anoutput of the pseudo-random sequence generator. The measurement devicealso includes a filter having an input coupled to an output of the ADC.

Also, the foregoing measurement device example embodiment may beimplemented to include one or more of the following additional features.The measurement device may also be implemented such that the ADCincludes a frequency divider coupled between an output of the firstoscillator and the filter input. In such an implementation, the filtermay be a low-pass filter. The measurement device may also be implementedsuch that the pseudo-random sequence generator further includes a LFSR.

The measurement device may also be implemented such that thepseudo-random sequence generator further includes a counter and a logicnetwork. The counter may include a counter reset input coupled to anoutput of a reference oscillator. The counter may also include a counterclock input coupled to the first oscillator output. In thisimplementation, the LFSR may include an enable input coupled to anoutput of the counter and an LFSR reset input coupled to the referenceoscillator output. The LFSR may also include an LFSR clock input coupledto the first oscillator output and an LFSR output coupled to the firstoscillator input. The logic network may include a first logic inputcoupled to the counter output, a second logic input coupled to the LFSRoutput, a first logic output coupled to the enable input of the LFSR,and a second logic output coupled to the first oscillator input.

The measurement device may also be implemented further to include asquare wave generator. The square wave generator may include an outputcoupled to an input of the sensor, and the sensor may include a MEMSelement having an underdamped response. The MEMS element may include afirst pressure-sensitive capacitor.

The measurement device may also be implemented further to include acapacitance bridge. The capacitance bridge may include a first bridgesection and a second bridge section. The first bridge section mayinclude the first pressure-sensitive capacitor and a first referencecapacitor coupled together at a first output node of the capacitancebridge. The second bridge section may include a secondpressure-sensitive capacitor and a second reference capacitor coupledtogether at a second output node of the capacitance bridge, such that acapacitance of the first pressure-sensitive capacitor varies withpressure more than a capacitance of the first reference capacitor. Inthis implementation, a capacitance of the second pressure-sensitivecapacitor varies with pressure more than a capacitance of the secondreference capacitor, and the first output node and the second outputnode of the capacitance bridge are coupled to the ADC.

Illustrative embodiments of the present invention have the advantage ofsuppressing narrow-band noise caused by resonance. An embodiment systemmay use, for example, a pseudo-random sampling clock jitter to increasethe width of the noise band so that it may be more easily filtered out.

Third Embodiment

In the third embodiment, the MEMS pressure sensor is implemented usingan array or MEMS pressure sensors that have varying dimensions, suchthat each of the MEMS pressure sensors resonate at differentfrequencies. Hence, when the MEMS pressure sensors are stimulated by theexcitation signal, the amplitude of the ringing may be reduced atvarious times due to the various resonant responses being out of phasewith each other. This is the case since the individual resonance signalsare added, for example, by connecting the sensors electrically inparallel. In order to reduce the ringing caused by an underdampedresponse of the MEMS pressure sensors are designed with an array of MEMScapacitive pressure sensors. Each capacitive pressure sensor is designedwith different dimensions such that the harmonic frequency for eachcapacitive sensor element is different than others in the array. Whenexcited with a square wave excitation signal, each capacitive sensorelement will ring with different resonant frequencies and attenuateharmonics that may stimulate the underdamped response of the MEMSpressure sensor.

FIGS. 12a show a schematic of a pressure sensitive MEMS array 1202 andFIG. 12b shows a corresponding layout configuration 1204 of the MEMSarray that includes twenty cells connected in parallel. Each unit cellis designed with different dimensions to have different resonantfrequency. For example, as shown in FIG. 12a , the first MEMS cell has adimension of 55.050 μm per side, the second MEMS cell has a dimension of55.500 μm per side and the last MEMS cell has a dimension of 63.975 μmper side for a total spread of +/−7.5%. It should be understood that theexample of FIGS. 12a and 12b is just one of many possible embodimentimplementations. In alternative embodiments of the present invention,the total spread may be different, as well as how the device sizes aredistributed.

Since each of the MEMS cells has a different dimension, each of the MEMScells has a different resonant frequency. Thus, each MEMS cell rings ata different frequency when stimulated by the input excitation signal. Atcertain time periods, the ringing amplitude may be small due todestructive interference, or the ringing amplitude may be larger due toconstructive interference. In various embodiments, ringing is reducedcompared to an assembly with an equally sized MEMS. In an embodiment,the sizes of the MEMS cells are chosen such that the resonantfrequencies of the MEMS cells are spread out such that the output of theMEMS cells may be measured and/or samples during suitable time periodsin which destructive interference occurs in order to reduce or minimizemeasurement error due to the ringing response of the MEMS cells.

In embodiments of the present invention, the total geometrical spreadbetween the smallest and the largest MEMS sensor cell, as well as howthe cell sizes are distributed may be determined by performingsimulations to determine how a size spread for a particular MEMS sensorcell effects a reduction in ringing for the composite response. In onespecific embodiment, the variation of the dimension of each cell islimited within 7.5%. However, this is just one example and inalternative embodiments of the present invention, the limit of variationof the dimension may be greater or less than 7.5%.

FIG. 13 shows a graph of output waveform utilizing a MEMS pressuresensor which includes each MEMS cell has the same dimension and sameresonant frequency. The horizontal axis represents the time inmilliseconds and vertical axis represents a normalized value of output.The wave form 1300 shows normalized amplitude of the output ringing whenexcited with a square wave input excitation signal. The waveformindicates the longer settling time and larger amplitude of the ringing.

FIG. 14 shows a graph of an output waveform that utilizes the MEMSpressure sensor having different dimensions for each unit cell andresonant frequencies spread out. The horizontal axis represents time inseconds and vertical axis represents normalized value of output. Asshown, the settling time of the time response is fast than theembodiment represented by the graph of FIG. 13, in which all MEMS cellshave the same dimension. In an embodiment, the ADC may sample the outputof the MEMS pressure sensor at times during which the ringing responseis at a minimum. In the specific embodiment represented by the graph ofFIG. 14, such times may include, for example, between about 2 μs andabout 4 μs or between 6 μs and 9 μs when the normalized ringing responseis less than 0.1%. In alternative embodiments, other settling times andsampling periods may be used depending on the particular embodiment andits specifications. Thus, in various embodiments, the MEMS resonancefrequency and the sampling clock frequency can vary over time ortemperature or supply voltage without losing the benefit of thisembodiment method.

FIG. 15 shows a graph of frequency spectrum of an output waveform wherethe resonant frequencies are spread out. The horizontal axis representsfrequency and vertical axis represents magnitude. In an embodiment, theresonant frequencies are spread out between 4.5 MHz to 6.2 MHz.

FIG. 16 shows a comparison graph that includes two simulated waveformsrepresenting the output voltage ringing. The waveform 1602 is of anembodiment that includes a MEMS pressure sensor array where each unitcell is having same dimension and resonant frequency, and waveform 1604is of an embodiment that includes a MEMS pressure sensor array whereeach unit cell has different dimension. The waveform 1602 shows ringingamplitude around 6 uV peak to peak, and waveform 1604 shows ringingamplitude around 1 uV peak to peak.

In accordance with various embodiments, circuits or systems may beconfigured to perform particular operations or actions by virtue ofhaving hardware, software, firmware, or a combination of them installedon the system that in operation causes or cause the system to performthe actions. One general aspect includes a method of performing ameasurement using a micro-electro-mechanical system (MEMS) sensorincluding MEMS sensors coupled in a bridge configuration, where aplurality of the MEMS sensors include a different resonant frequencies,the method including: applying an excitation signal to a first port ofthe bridge configuration, where each of the plurality of the MEMSsensors is stimulated by the excitation signal; measuring a signal at asecond port of the bridge configuration; and determining a measuredvalue based on the measuring the signal. Other embodiments of thisaspect include corresponding circuits and systems configured to performthe various actions of the methods.

Implementations may include one or more of the following features. Themethod where the MEMS sensor includes a MEMS pressure sensor and themeasured value includes a pressure. The method where the bridgeconfiguration includes a first branch having a first group of the MEMSsensors coupled to a first capacitor, and a second branch having asecond group of MEMS sensors coupled to a second capacitor. The methodwherein each of the plurality of the MEMS sensors includes differentsize dimensions. The method where the size dimensions vary by about+/−7.5%. The method where the size dimensions are evenly distributed.The method where measuring a signal at a second port of the bridgeconfiguration includes performing an A/D conversion. The method where atransient response of the bridge configuration includes ringing at thedifferent resonant frequencies, and the ringing includes time intervalsof constructive interference and intervals of destructive interference.The method where measuring a signal at a second port of the bridgeconfiguration includes measuring the signal at the second port of thebridge configuration during an interval of destructive interference. Themethod where measuring the signal further includes sampling the signalduring the interval of destructive interference. The method wheremeasuring the signal further includes performing an A/D conversion ofthe signal during the interval of destructive interference.Implementations of the described techniques may include hardware, amethod or process, or computer software on a computer-accessible medium.

One general aspect includes a system including: amicro-electro-mechanical system (MEMS) sensor array including a bridge,the bridge including a first bridge section and a second bridge section,where the first bridge section includes a first pressure sensitive MEMSsensor coupled to a first reference MEMS capacitor, where the firstpressure sensitive MEMS sensor includes a first array of multiple MEMSsensors having different resonant frequencies. Other embodiments of thisaspect include corresponding circuits and systems configured to performthe various actions of the methods.

Implementations may include one or more of the following features. Thesystem where the second bridge section includes a second pressuresensitive MEMS sensor coupled to a second reference MEMS capacitor,where the second pressure sensitive MEMS sensor includes a second arrayof multiple MEMS sensors having different resonant frequencies. Thesystem where: the multiple MEMS sensors of the first array are coupledin parallel; and the multiple MEMS sensors of the second array arecoupled in parallel. The system where the multiple MEMS sensors of thefirst array are rectangular. The system where multiple MEMS sensors ofthe first array each have different dimensions. The system where thedifferent dimensions include different lengths. The system where thedifferent lengths have a variation of about +/−7.5%. The system furtherincluding: an excitation generator having an output coupled to a firstport of the MEMS sensor array; and a measurement circuit having an inputcoupled to a second port of the MEMS sensor array. The system where themeasurement circuit includes an A/D converter. The system furtherincluding a filter coupled to an output of the A/D converter. The systemwhere the filter includes a low pass filter. Implementations of thedescribed techniques may include hardware, a method or process, orcomputer software on a computer-accessible medium.

An advantage of some embodiments includes the ability to reduce theeffect of ringing when capacitive MEMS array is designed with MEMS cellshaving different dimensions and resonant frequencies.

Fourth Embodiment

In a fourth embodiment, an oversampled analog to digital converter (ADC)is used to monitor the output of the MEMS sensor. In order to alleviateidle tones in the oversampled ADC, a dithered clock is used to operatethe oversampled ADC. In some embodiments, the dither clock signal may begenerated according to the second embodiment disclosed herein.

According to various embodiments, the conversion of the analog signalinto the digital domain is performed using a sigma-delta analog todigital converter (ADC). Various embodiment sigma-delta ADCs includefeedback and reference voltage supplies. For pressure sensing, themeasured signal is generally at very low frequencies near DC. Forexample, pressure sensing may measure input signals from 0 to 10 Hz. Theinventors have determined that idle tones present in the sigma-delta ADCinteract with noise in the reference voltage supply in a multiplicativemanner to produce an error component at DC. In various embodiments, thesigma-delta ADC is provided with a dithered clock in order to spread thenoise components and reduce or remove the error component at DC. In suchembodiments, the dithered clock is used as a system clock for theinterface circuit including, for example, the voltage supply circuit,the sigma-delta ADC, an output filter, or other components.

FIG. 17 illustrates a system block diagram of an embodiment MEMSpressure sensor system 1700 including MEMS pressure sensor 1702, outputcircuit 1704, dithered clock 1706, and supply circuit 1708. According tovarious embodiments, MEMS pressure sensor 1702 transduces physicalpressure measurement P_(MES) into analog signal A_(MES). MEMS pressuresensor 1702 receives a voltage reference V_(REF) from supply circuit1708, which may generate a switching voltage as voltage referenceV_(REF) based on dithered clock signal CLK from dithered clock 1706.Supply circuit 1708 may also provide voltage reference V_(REF) to outputcircuit 1704. Output circuit 1704 receives analog signal A_(MES) andgenerates digital pressure signal D_(MES) based on analog signalA_(MES).

In various embodiments, output circuit 1704 operates to amplify,convert, and filter the analog signal A_(MES) in order to generatedigital pressure signal D_(MES). In such embodiments, output circuitincludes a sigma-delta ADC that converts analog signal A_(MES) intodigital pressure signal D_(MES) based on a sampling time controlled bydithered clock signal CLK from dithered clock 1706. The operation of thesigma-delta ADC based on a dithered clock may reduce or remove DC noisecomponents generated through an interaction of idle tones from thesigma-delta ADC and noise in voltage reference V_(REF).

FIG. 18 illustrates a schematic block diagram of a further embodimentMEMS pressure sensor system 1800 including differential outputcapacitive MEMS pressure transducer 1802 and application specificintegrated circuit (ASIC) 1803, which further includes output circuit1804, dithered clock 1806, and voltage reference supply 1808. Outputcircuit 1804 includes amplifier 1810, incremental sigma-delta ADC 1812,and decimation filter 1814. MEMS pressure sensor system 1800 may be oneimplementation of MEMS pressure sensor system 1700 described hereinabovein reference to FIG. 17.

According to various embodiments, differential output capacitive MEMSpressure transducer 1802 transduces physical pressure signals into adifferential analog output including analog signals A+ and A−.Differential output capacitive MEMS pressure transducer 1802 includesvariable capacitance structure 1820 and variable capacitance structure1824 connected with reference capacitive structure 1822 and referencecapacitive structure 1826 as a capacitive bridge with analog signals A+and A− output from center nodes of each branch of the capacitive bridgeas shown. In such embodiments, reference capacitive structure 1822 andreference capacitive structure 1826 may be formed of electricallyconductive structures, i.e., forming parallel plates, separated by adielectric spacer where the spacing of the electrically conductivestructures is fixed and does not vary in response to pressure changes.Variable capacitance structure 1820 and variable capacitance structure1824 are formed of electrically conductive structures separated by aspacing distance where the spacing of the electrically structures isdependent on the pressure applied to the electrically conductivestructures. For example, variable capacitance structure 1820 andvariable capacitance structure 1824 may each include a deflectablemembrane formed over a sealed cavity above a substrate with anelectrically conductive diffusion region below the membrane. In suchembodiments, the membrane of the variable capacitance structure maydeflect due to a pressure difference between the external surface andthe sealed cavity. Such deflections affect the capacitance between themembrane and the electrically conductive diffusion region, which ismeasured at electrical contacts to the membrane and the electricallyconductive diffusion region. Reference capacitive structure 1822 andreference capacitive structure 1826 may each have a similar structurewhere the cavity is filled with the dielectric spacer material. In otherembodiments, many types of capacitive pressure sensors may be used fordifferential output capacitive MEMS pressure transducer 1802 includingcapacitive comb drive structures, multiple plate released capacitiveplate structures, or other capacitive MEMS structures, for example.

In various embodiments, the differential analog output including analogsignals A+ and A− is supplied from differential output capacitive MEMSpressure transducer 1802 to amplifier 1810, which amplifies thedifferential signal and provides the amplified analog electrical signalproportional to the measured physical pressure to incrementalsigma-delta ADC 1812. In other embodiments, incremental sigma-delta ADC1812 may be any type of sigma-delta ADC. In one particular embodiment,incremental sigma-delta ADC 1812 operates for a set duration or numberof samples before ending operation and is thus referred to asincremental. Such an embodiment may reduce power consumption.Incremental sigma-delta ADC 1812 begins operation, e.g., wakeup, on afixed time delay or in response to a pressure change above a thresholdlevel. In some embodiments, ADC 1812 is powered up for a certain periodof time as determined, for example by a target accuracy setting, andthen turned off until a next conversion is requested.

In various embodiments, incremental sigma-delta ADC 1812 operatesaccording to dithered clock signal CLK to generate a digital outputsignal proportional to the input amplified analog electrical signal thatis proportional to the measured physical pressure from differentialoutput capacitive MEMS pressure transducer 1802. Incremental sigma-deltaADC 1812 includes a feedback mechanism that continually adjusts thedigital output signal. Further description of two example sigma-deltaADCs is provided herein below in reference to FIGS. 21a and 21 b.

In various embodiments, the digital output signal from incrementalsigma-delta ADC 1812 may have a high bit rate. Incremental sigma-deltaADC 1812 may include a sampling rate, i.e., an over-sampling rate, thatis on the order of 1000 or 10,000 times higher than the intendedsampling rate, for example. In one specific embodiment, incrementalsigma-delta ADC 1812 may output the digital signal based on a 160 kHzsampling rate, which corresponds to 160,000 samples per second. For suchas system, the intended digital output signal may be only 10 Hz. In suchembodiments, decimation filter 1814 reduces the 160 kHz signal down to a10 Hz and outputs digital output signal D_(OUT), which is proportionalto the measured physical pressure signal from differential outputcapacitive MEMS pressure transducer 1802, at the 10 Hz frequency. Thus,decimation filter 1814 reduces the bit rate by a factor of 16,000. Inother embodiments, decimation filter 1814 may reduce the bit rate byother factors.

According to various embodiments, dithered clock 1806 supplies ditheredclock signal CLK to incremental sigma-delta ADC 1812 for controlling thesampling rate of the sigma-delta ADC. In various embodiments, ditheredclock signal CLK may also be provided to voltage reference supply 1808,amplifier 1810, or decimation filter 1814. Dithered clock 1806 generatesdithered clock signal CLK with jitter or random periods. Generally, aclock signal is generated with a fixed or constant period, including,for example, constant rising or logic high durations and constantfalling or logic low durations. In the case of dithered clock 1806, therising or logic high durations and falling or logic low durations areadjusted. In such embodiments, the adjustments of the dithered clocksignal CLK may be random or pseudo-random. Thus, dithered clock signalCLK is generated to intentionally include substantial clock jitter withvaried rising or logic high durations or varied falling or logic lowdurations.

In various embodiments, voltage reference supply 1808 supplies voltagereference V_(REF) to incremental sigma-delta ADC 1812 in order to supplypower to the ADC. Voltage reference supply 1808 may also providereference voltages to differential output capacitive MEMS pressuretransducer 1802 in order to bias the capacitive structures.Specifically, voltage reference supply 1808 provides positive referencevoltage V+ and negative reference voltage V−. In some particularembodiments, voltage reference supply 1808 provides pulsed referencevoltages to differential output capacitive MEMS pressure transducer1802. In such embodiments, voltage reference supply 1808 may include achopper switch to switch the reference voltages supplied to differentialoutput capacitive MEMS pressure transducer 1802.

In various embodiments, differential output capacitive MEMS pressuretransducer 1802 and ASIC 1803 are formed on separate wafers or dies. Inother embodiments, differential output capacitive MEMS pressuretransducer 1802 and ASIC 1803 are formed on a same wafer or die, such asa single integrated circuit (IC) die.

FIGS. 19a and 19b illustrate waveform diagrams of example noise signalsgenerated in a sigma-delta ADC. FIG. 19a shows plot 1901 depicting afast Fourier transform (FFT) of the output of the sigma-delta ADC, whichalso shows idle tone 1912 and idle tone 1914 at about 65 kHz and 95 kHz,respectively. Idle tones are unwanted bit sequences at specificfrequencies, as shown, in the output that are not based on the inputsignal to the sigma-delta ADC. These idle tones are an artifact of thefeedback mechanism in the sigma-delta ADC, and may be present when theinput signal to the ADC is not “busy.” In some cases, the idle tones maybe especially strong when the ADC uses a one-bit (2-level) quantizer.Such a situation is applicable to embodiment pressure sensors that use asigma-delta ADC with a one-bit quantizer for good linearity.

FIG. 19b shows plot 1900 depicting the number of bits, which indicatesthe approximate accuracy, of the sigma-delta ADC. Plot 1900 wasgenerated by introducing an unwanted sinewave at the sigma-delta ADC'sreference voltage in order to determine the robustness of thesigma-delta ADC with respect to supply voltage ripple. The frequency ofthis sinewave signal is swept from 0 to 160 kHz and the integrated noisewas measured. Accordingly, the ADC is most sensitive to disturbers atVref which have the same frequency as the idle tones. Since idle tonefrequency varies with the DC-input level of the sigma-delta ADC, whichin some embodiments may be measured pressure from a MEMS sensor, thesensitivity to disturbers changes with the ADC input. Such a behavior isproblematic in some embodiments because it is difficult to predict ifthe idle tone behavior leads to a problem in a particular application.In this example sigma-delta ADC, there is a sharp decrease in accuracyat about 65 kHz and 95 kHz at points 1902 and 1904, respectively thatcorrespond to the frequency of idle tones 1912 and 1914 shown in FIG. 19a.

As described briefly hereinabove, the inventors have determined thatidle tones present in a sigma-delta ADC interact with noise in thereference voltage supply in a multiplicative manner to produce an errorcomponent at DC. Thus, idle tone 1902 and idle tone 1904 may interactwith noise in the reference voltage supplied to the sigma-delta ADC. Asdescribed hereinabove in reference to FIGS. 17 and 18, variousembodiments include providing dithered clock signal CLK to thesigma-delta ADC. In such embodiments, the sharp value of the idle tones,such as idle tone 1902 and idle tone 1904, is dispersed across frequencyspectrum and the noise component at DC produced from the idle tone incombination with the noise in the reference voltage supply is reduced orremoved. An embodiment noise plot is shown in FIG. 20.

FIG. 20 illustrates a waveform diagram of noise signals generated in asigma-delta ADC without a dithered clock (plot 2000) and with a ditheredclock (plot 2001) showing a comparison of resolution with respect toreference voltage disturber frequency. As shown, the example sigma-deltaADC with a standard non-dithered clock (plot 2000) indicates a loss inperformance at points 2002 and 2004 corresponding to idle tones 1912 and1914 shown in FIG. 19a above. However, the embodiment sigma-delta ADCwith a dithered clock does not substantially loose noise performance atidle tone frequencies. As shown in plot 2001, there is a substantiallyreduced loss of noise performance at the 65 kHz and 95 kHz idle tonefrequencies. In such embodiments, the clock dithering may spread thefrequency spectrum of idle tones and reduce or remove the DC noisecomponent produced by the multiplicative interaction of the idle tonesand the noise in the reference voltage supply.

FIGS. 21a and 21b illustrate schematic block diagrams of embodimentsigma-delta ADCs 2100 and 2101. FIG. 21a shows discrete time sigma-deltaADC 2100 including sampling switch 2102, loop filter 2104, comparator2106, digital to analog converter (DAC) 2108, and adder 2110. Accordingto various embodiments, discrete time sigma-delta ADC 2100 receivesanalog input signal A_(IN) at sampling switch 2102. Analog input signalA_(IN) may be an amplified analog signal received from an amplifier thatis coupled to a capacitive MEMS pressure transducer, such as fromamplifier 1810 described hereinabove in reference to FIG. 18. Thus,analog input signal A_(IN) may be proportional to a measured physicalpressure signal, for example.

In various embodiments, sampling switch 2102 is controlled by ditheredclock signal CLK, which may be provided from dithered clocks 1706 or1806 as described hereinabove in reference to FIGS. 17 and 18. Ditheredclock signal CLK causes sampling switch 2102 to open and close accordingto the sampling rate equal to the frequency of dithered clock signalCLK. Thus, sampling switch 2102 generates sampled analog input signalSA_(IN), which is provided through adder 2110 to loop filter 2104. Byvirtue of sampling, the analog signal is no longer a continuous signal,but is instead a discretely sampled analog input signal SA_(IN). In suchembodiments, loop filter 2104 may be implemented as a low pass filter(LPF) in order to remove higher frequency components. In someembodiments, loop filter 2104 is implemented as an integrator.

According to various embodiments, after filtering in loop filter 2104,the sampled and filtered analog input signal is provided to comparator2106, which compares the input signal to a threshold value. For example,the threshold value may be 0 V. Based on the comparison, comparator 2106provides digital output signal D_(OUT). The stream of bits in digitaloutput signal D_(OUT) is proportional to analog input signal A_(IN).Further, digital output signal D_(OUT) is provided through DAC 2108 backto adder 2110. In such embodiments, DAC 2108 is supplied by voltagereference V_(REF) such as from supply circuit 1708 or voltage referencesupply 1808 described hereinabove in reference to FIGS. 17 and 18.

As discussed herein, the feedback loop of some sigma-delta ADCs maygenerate idle tones and there may be noise in voltage reference V_(REF).In such ADCs, these two error sources may be multiplicatively combinedby a DAC to form a DC error component. In various embodiments, theintroduction of dithered clock signal CLK from a dithered clock spreadsthe frequency of the idle tones and reduces or removes the DC errorcomponent. Adder 2110 combines the reconverted analog output of DAC 2108with sampled analog input SA_(IN) in order to provide feedback forimproved performance.

FIG. 21b shows continuous time sigma-delta ADC 2101 including loopfilter 2105, clocked comparator 2112, DAC 2108, and adder 2110.According to various embodiments, continuous time sigma-delta ADC 2101operates as described hereinabove in reference to discrete timesigma-delta ADC 2100 in FIG. 21a , where sampling switch 2102 isremoved, loop filter 2104 is replaced with loop filter 2105, andcomparator 2106 is replaced with clocked comparator 2112. In suchembodiments, analog input signal A_(IN) is provided through adder 2110to loop filter 2105. Loop filter 2105 may operate as described inreference to loop filter 2104, but is arranged to receive a continuoustime signal in analog input signal A_(IN) instead of a discretelysampled signal in sampled analog input signal SA_(IN).

In various embodiments, clocked comparator 2112 compares the filteredanalog input signal to a threshold voltage and provides the result ofthe conversion with the dithered clock signal CLK to generate digitaloutput signal D_(OUT). In some embodiments, the threshold voltage may bezero volts, VDD/2 and/or other threshold voltages. In such embodiments,dithered clock signal CLK determines the sampling rate of continuoustime sigma-delta ADC 2101. As described hereinabove in reference to FIG.21a , DAC 2108 provides feedback through adder 2110.

In such embodiments, providing dithered clock signal CLK to clockedcomparator 2112 provides the same benefits as described hereinabove inreference to dithered clock signal CLK in the other Figures.

FIG. 22 illustrates a block diagram of an embodiment method of operation2200 for a sensor. Method of operation 2200 includes steps 2202-2212.According to various embodiments, step 2202 includes transducing apressure signal into an electrical signal. The pressure signal may bemeasured and transduced using a capacitive MEMS pressure transducer.Step 2204 includes generating an amplified electrical signal byamplifying the electrical signal. For example, the electrical signal maybe amplified by a differential input amplifier. Step 2206 includesgenerating a dithered clock signal. In such embodiments, a ditheredclock is included in the sensor system to generate the dithered clocksignal. In step 2207, a reference voltage is provided to the sigma-deltaADC.

According to various embodiments, step 2208 includes converting theamplified electrical signal into a digital signal using a sigma-deltaADC operating with a sampling time controlled by the dithered clocksignal generated in step 2206. In various embodiments, steps 2202-2208may be rearranged and performed in other orders and method of operation2200 may be modified to include additional steps.

In accordance with various embodiments, circuits or systems may beconfigured to perform particular operations or actions by virtue ofhaving hardware, software, firmware, or a combination of them installedon the system that in operation causes or cause the system to performthe actions. One general aspect includes a sensor including: amicroelectromechanical systems (MEMS) pressure transducer; an amplifiercoupled to the MEMS pressure transducer; a sigma-delta analog to digitalconverter (ADC) coupled to the amplifier; a dithered clock coupled tothe sigma-delta ADC and configured to control a sampling time of thesigma-delta ADC using a dithered clock signal; and a supply voltagecircuit coupled to the sigma-delta ADC and the dithered clock, where thesupply voltage circuit is configured to operate based on the ditheredclock signal. Other embodiments of this aspect include correspondingcircuits and systems configured to perform the various actions of themethods.

One general aspect includes a method of operating a sensor, the methodincluding: transducing a pressure signal into an electrical signal;generating an amplified electrical signal by amplifying the electricalsignal; generating a dithered clock signal; converting the amplifiedelectrical signal into a digital signal using a sigma-delta analog todigital converter (ADC) operating with a sampling time controlled by thedithered clock signal; generating a reference voltage based on thedithered clock signal; and providing the reference voltage to thesigma-delta ADC. Other embodiments of this aspect include correspondingcircuits and systems configured to perform the various actions of themethods.

One general aspect includes a microelectromechanical systems (MEMS)capacitive pressure sensor system including: a differential output MEMScapacitive pressure sensor including: a first reference capacitivestructure, a first variable capacitance structure configured to vary afirst capacitance value in reference to a first pressure signal, a firstoutput coupled between the first reference capacitive structure and thefirst variable capacitance structure, a second reference capacitivestructure, a second variable capacitance structure configured to vary asecond capacitance value in reference to a second pressure signal, and asecond output coupled between the second reference capacitive structureand the second variable capacitance structure; a differential amplifiercoupled to the first output and the second output of the differentialoutput MEMS capacitive pressure sensor; a sigma-delta analog to digitalconverter (ADC) coupled to an output of the differential amplifier; adithered clock coupled to the sigma-delta ADC and configured to controla sampling time of the sigma-delta ADC using a dithered clock signal;and a supply voltage circuit coupled to the sigma-delta ADC. Otherembodiments of this aspect include corresponding circuits and systemsconfigured to perform the various actions of the methods.

In some specific embodiments, a MEMS pressure transducer with asigma-delta ADC operated according to a dithered clock signal generatedby a dithered clock is particularly advantageous. In such specificembodiments, the absolute pressure measurement or very low frequencypressure measurement is particularly affected by DC noise from idletones and reference voltage supply noise as described hereinabove. Thus,such specific embodiments include particular advantages of decreasednoise or reduced error components at DC and very low frequencymeasurements, which may allow improved sensitivity or greaterresolution.

A further advantage of some embodiments includes having a more robustsensor that is less susceptible to disturbers at sensor supply nodes,especially tonal disturbers having a same or similar frequency as ADCidle tones.

While this invention has been described with reference to illustrativeembodiments, this description is not intended to be construed in alimiting sense. Various modifications and combinations of theillustrative embodiments, as well as other embodiments of the invention,will be apparent to persons skilled in the art upon reference to thedescription. It is therefore intended that the appended claims encompassany such modifications or embodiments.

What is claimed is:
 1. A measurement method, comprising: generating, bya sensor, a response signal in response to an excitation signal;generating a sampling clock signal in accordance with a pseudo-randomjitter; sampling the response signal in accordance with the samplingclock signal to determine a plurality of digital samples; and combiningthe plurality of digital samples to form a measurement sample.
 2. Themethod of claim 1, wherein: the generating the sampling clock signalcomprises generating a variable clock signal having a variable clockfrequency that switches in accordance with a switching frequency; and aperiod of the sampling clock signal is an integer multiple of a periodof the variable clock signal.
 3. The method of claim 2, wherein: thesensor has an underdamped response; the switching frequency is not lessthan 0.9 times a mechanical resonant frequency of the sensor; and theswitching frequency is not greater than 1.1 times the mechanicalresonant frequency of the sensor.
 4. The method of claim 2, wherein thegenerating the variable clock signal comprises: generating, by a linearfeedback shift register (LFSR), a LFSR state signal in accordance withthe variable clock signal and a reference oscillator signal.
 5. Themethod of claim 4, wherein: the generating the variable clock signalfurther comprises: generating a counter signal in accordance with thevariable clock signal and the reference oscillator signal; generating anLFSR enable signal in accordance with the counter signal; and generatinga frequency select signal in accordance with the LFSR state signal andthe counter signal; the generating the LFSR state signal is further inaccordance with the LFSR enable signal; and the generating the variableclock signal is further in accordance with the frequency select signal.6. The method of claim 1, wherein: the excitation signal comprises asquare wave; the sensor comprises a micro-electro-mechanical systems(MEMS) element; and the MEMS element comprises a firstpressure-sensitive capacitor.
 7. The method of claim 6, wherein: thesensor further comprises a capacitance bridge; the capacitance bridgecomprises a pair of bridge sections, each respectively comprising apressure-sensitive capacitor and a reference capacitor coupled togetherat an output node; a capacitance of the pressure-sensitive capacitorvaries with pressure more than a capacitance of the reference capacitor;and the response signal comprises output signals of the pair of bridgesections.
 8. The method of claim 7, wherein: the capacitance of thereference capacitor changes with temperature in a known relationshiprelative to a change with temperature of the capacitance of thepressure-sensitive capacitor.
 9. The method of claim 1, wherein thecombining the plurality of digital samples comprises at least one ofaveraging first digital samples of the plurality of digital samples orcalculating a median value of the first digital samples.
 10. Ameasurement circuit comprising a sensor, wherein the measurement circuitis configured to: generate a response signal in response to anexcitation signal; generate a sampling clock signal in accordance with apseudo-random jitter; sample the response signal in accordance with thesampling clock signal to determine a plurality of digital samples; andcombine the plurality of digital samples to form a measurement sample.11. The measurement circuit of claim 10, further configured to: generatea variable clock signal having a variable clock frequency that switchesin accordance with a switching frequency, wherein a period of thesampling clock signal is an integer multiple of a period of the variableclock signal.
 12. The measurement circuit of claim 11, wherein: thesensor has an underdamped response; the switching frequency is not lessthan 0.9 times a mechanical resonant frequency of the sensor; and theswitching frequency is not greater than 1.1 times the mechanicalresonant frequency of the sensor.
 13. The measurement circuit of claim11, further comprising a linear feedback shift register (LFSR)configured to generate a LFSR state signal in accordance with thevariable clock signal and a reference oscillator signal.
 14. Themeasurement circuit of claim 13, wherein: the measurement circuit isfurther configured to: generate a counter signal in accordance with thevariable clock signal and the reference oscillator signal; generate thevariable clock signal in accordance with a frequency select signal;generate an LFSR enable signal in accordance with the counter signal;and generate the frequency select signal in accordance with the LFSRstate signal and the counter signal; and the LFSR is further configuredto generate the LFSR state signal in accordance with the LFSR enablesignal.
 15. The measurement circuit of claim 10, wherein: the excitationsignal comprises a square wave; the sensor comprises amicro-electro-mechanical systems (MEMS) element; and the MEMS elementcomprises a first pressure-sensitive capacitor.
 16. The measurementcircuit of claim 15, wherein: the sensor further comprises a capacitancebridge; the capacitance bridge comprises a pair of bridge sections, eachrespectively comprising a pressure-sensitive capacitor and a referencecapacitor coupled together at an output node; a capacitance of thepressure-sensitive capacitor varies with pressure more than acapacitance of the reference capacitor; and the response signalcomprises output signals of the pair of bridge sections.
 17. Themeasurement circuit of claim 16, wherein the capacitance of thereference capacitor changes with temperature in a known relationshiprelative to a change with temperature of the capacitance of thepressure-sensitive capacitor.
 18. The measurement circuit of claim 10,wherein the measurement sample comprises at least one of an average offirst digital samples of the plurality of digital samples or a medianvalue of the first digital samples.
 19. A measurement device,comprising: a sensor; an analog-to-digital converter (ADC) coupled to anoutput of the sensor, the ADC comprising: a pseudo-random sequencegenerator; and a first oscillator comprising an input coupled to anoutput of the pseudo-random sequence generator; and a filter comprisingan input coupled to an output of the ADC.
 20. The measurement device ofclaim 19, wherein: the ADC comprises a frequency divider coupled betweenan output of the first oscillator and the filter input; and the filtercomprises a low-pass filter.
 21. The measurement device of claim 20,wherein the pseudo-random sequence generator further comprises a linearfeedback shift register (LFSR).
 22. The measurement device of claim 21,wherein: the pseudo-random sequence generator further comprises acounter and a logic network; the counter comprises: a counter resetinput coupled to an output of a reference oscillator; and a counterclock input coupled to the first oscillator output; the LFSR comprises:an enable input coupled to an output of the counter; an LFSR reset inputcoupled to the reference oscillator output; an LFSR clock input coupledto the first oscillator output; and an LFSR output coupled to the firstoscillator input; and the logic network comprises: a first logic inputcoupled to the counter output; a second logic input coupled to the LFSRoutput; a first logic output coupled to the enable input of the LFSR;and a second logic output coupled to the first oscillator input.
 23. Themeasurement device of claim 19, further comprising a square wavegenerator, wherein: the square wave generator comprises an outputcoupled to an input of the sensor; the sensor comprises amicro-electro-mechanical systems (MEMS) element having an underdampedresponse; and the MEMS element comprises a first pressure-sensitivecapacitor.
 24. The measurement device of claim 23, further comprising acapacitance bridge, wherein: the capacitance bridge comprises a firstbridge section and a second bridge section; the first bridge sectioncomprises the first pressure-sensitive capacitor and a first referencecapacitor coupled together at a first output node of the capacitancebridge; and the second bridge section comprises a secondpressure-sensitive capacitor and a second reference capacitor coupledtogether at a second output node of the capacitance bridge, wherein: acapacitance of the first pressure-sensitive capacitor varies withpressure more than a capacitance of the first reference capacitor; acapacitance of the second pressure-sensitive capacitor varies withpressure more than a capacitance of the second reference capacitor; andthe first output node and the second output node of the capacitancebridge are coupled to the ADC.
 25. The measurement device of claim 24,wherein the measurement device occupies a volume that is not greaterthan 10 cubic millimeters.